Channel estimation in a multi carrier transmit diversity system

ABSTRACT

A method and a circuit ( 44 ) for estimating channel coefficients in a multi carrier transmit diversity system operating in accordance with a block-coding scheme is described. The method comprises determining from a receive signal (Y) for each channel estimated channel coefficients (ĥ) comprising interference components from adjacent channels artificially introduced during the estimation, deriving estimates for interference components and determining interference-compensated estimates (Ĥ F+IC ) for the channel coefficients on the basis of the estimates for the interference components.

BACKGROUND OF THE INVENTION

[0001] 1. Technical Field

[0002] The present invention relates to the field of transmit antennadiversity and in particular to a method of estimating channelcoefficients in a multi carrier transmit diversity system. The inventionalso relates to an estimating circuit for performing channel estimationoperations and to a transceiver of a wireless communication systemcomprising such an estimating circuit.

[0003] 2. Discussion of the Prior Art

[0004] Peak transmission rates in wireless communication systems havesteadily increased during the last years. However, peak transmissionrates are still limited for example due to path loss, limited spectrumavailability and fading.

[0005] Transmitter diversity is a highly effective technique forcombating fading in wireless communications systems. Several differenttransmit diversity schemes have been proposed. In Li, Y.; Chuang, J. C.;Sollenberger, N. R.: Transmitter diversity for OFDM systems and itsimpact on high-rate data wireless networks, IEEE Journal on Selec.Areas, Vol. 17, No. 7, July 1999 the transmit diversity schemes ofdelay, permutation and space-time coding are exemplarily described.According to the delay approach, a signal is transmitted from a firsttransmitter antenna and signals transmitted from further transmitterantennas are delayed versions of the signal transmitted from the firsttransmitter antenna. In the permutation scheme, the modulated signal istransmitted from a first transmitter antenna and permutations of themodulated signal are transmitted from further transmitter antennas.Thus, the signal transmitted from the transmitter antennas can bederived from a matrix composed of data words in the form of themodulated signal and of permutations of the modulated signal. By meansof space-time coding a signal is encoded into several data words andeach data word is transmitted from a different transmitter antenna.During transmission the data words are spread (i.e., multiplexed) in thetime domain by successively transmitting the data symbols of each dataword over a single carrier frequency.

[0006] A further transmit diversity scheme for a multicarrier system isspace-frequency coding. By means of space-frequency coding a signal isencoded into several data words and each data word is spread (i.e.,multiplexed) in the frequency domain by transmitting the data symbols ofeach data word on orthogonal frequencies, i.e. orthogonal subcarriers.An exemplary scheme for space-frequency coding is described in Mudulodu,S.; Paulraj, A.: A transmit diversity scheme for frequency selectivefading channels, Proc. Globecom, San Francisco, pp. 1089-1093, November2000. According to the multicarrier system described in this paper, thedata words relating to an encoded signal are preferably multiplexed inthe time domain although orthogonal frequencies are available and thedata words could thus also be multiplexed in the frequency domain. Thisis due to the fact that if multiplexing in the frequency domain wasutilized the employed frequencies, i.e. subcarriers, had to see the samechannel, which may not always be possible in a frequency selectivefading channel. However, in case the subcarriers experience the samechannel, it is stated that either multiplexing in the time domain ormultiplexing in the frequency domain or a combination of the two may beused. By combining multiplexing in the time domain and in the frequencydomain the data symbols of a data word are simultaneously multiplexed inthe time domain and in the frequency domain. This means that the dataword is spread both across time and across frequencies.

[0007] Another transmit diversity scheme is described in U.S. Pat. No.6,088,408. According to this transmit diversity scheme data are codedand transmitted as individual data blocks. Each data block comprisesseveral data words and each data word contains data symbols derived froman input data signal. During transmission of the data blocks, theindividual data words are spread in the time domain. Therefore, thetransmit diversity scheme described in U.S. Pat. No. 6,088,408 can bereferred to as space-time block-coding (STBC). The main features of STBCare that each data symbol is transmitted from each transmit antenna andthat the antenna signals of different transmit antennas are orthogonalto each other. Orthogonal STBC data blocks can be designed for anarbitrary number of transmit antennas.

[0008] An important feature on a receiving side of a multi carriertransmit diversity system is a characterization of the individualtransmit channels. Consequently, a channel estimation has to beperformed as described in Li, Y.; Chuang, J. C.; Sollenberger, N. R.:Transmitter diversity for OFDM systems and its impact on high-rate datawireless networks, IEEE Journal on Selec. Areas, Vol. 17, No. 7, July1999 and in U.S. Pat. No. 6,088,408.

[0009] Departing from the various channel estimation approaches known inthe art there is a need for a more accurate method of estimating channelcoefficients in a multi carrier transmit diversity system operating inaccordance with a block-coding scheme. There is also a need for anestimating circuit for performing the corresponding estimating methodand for a transceiver comprising such an estimating circuit.

BRIEF DESCRIPTION OF THE INVENTION

[0010] The existing need is satisfied by a method of estimating channelcoefficients in a multi carrier transmit diversity system operating inaccordance with a block-coding scheme, comprising determining from areceive signal for each channel individually estimated channelcoefficients comprising artificially introduced interference componentsfrom adjacent channels, deriving estimates for the interferencecomponents and determining interference-compensated estimates for thechannel coefficients utilizing the estimates for the interferencecomponents.

[0011] The channel estimation method of the invention is not restrictedto a specific block-coding scheme as long as the utilized block-codingscheme enables to generate from a data signal block code matrices,preferably in the form of data blocks comprising data words, whereineach data word contains data symbols derived from the data signal. Forexample, the transmit diversity schemes of space-time block-coding, ofspace-frequency block-coding (SFBC) and of permutation in the timedomain or in the frequency domain allow to generate such data blocks.Preferably, the generated data blocks have the structure of a matrixsimilar to an STBC or SFBC code matrix.

[0012] The channel estimation method according to the invention does notrequire that the transmit diversity scheme guarantees full transmitdiversity and orthogonality. For example, the invention does notnecessitate that each information symbol comprised within the datasignal is transmitted from each transmitter antenna. Nonetheless, apreferred embodiment of the invention comprises the feature of fulltransmit diversity and orthogonality.

[0013] Moreover, the invention is not restricted to any number oftransmit and receive antennas. Preferably, the number of data words perdata block equals the number of transmit antennas such that each dataword of a data block may be transmitted from an individual transmitterantenna. If more than one receive antenna is provided, the receivediversity scheme of maximum-ratio combining can be applied. However,other receive diversity schemes may be used as well.

[0014] Various alternatives for determining an estimate for the channelcoefficients exist. For example, the determination of the estimatedchannel coefficients can be based on the assumption that the channels donot change during a specific amount of instants required to transmit twoor more data symbols or a complete data word comprising N data symbols.In the present case, the expression instants denotes either specificmoments in time or specific frequencies, depending on whether the datawords are spread in time direction or in frequency direction. This meansthat if the data words are spread in time direction, the specific amountof instants corresponds to a specific time interval. On the other hand,if the data words are spread in the frequency direction, the specificamount of instants corresponds to a specific frequency band.

[0015] The above assumption or other assumptions regarding the channelcoefficients may have the consequence that the channel coefficientsestimated for two or more adjacent instants are identical. Thus, in agraphical representation the estimated channel coefficients for aspecific channel will change in a stepwise manner in the time orfrequency direction (FIG. 11). The width of the steps corresponds to theamount of instants during which it is assumed that the channels do notchange. If, for example, it is assumed that the channels do not changeduring N instants required to transmit a data word comprising N datasymbols, the steps in a graphical representation of the estimatedchannel coefficients will have a width of N instants.

[0016] The estimated channel coefficients can be derived from theproduct of a known data block comprised within the receive signal andthe Hermitian of this known data block. The known data block comprisedwithin the receive signal may be a standardized signal portion like apreamble of the data signal to be transmitted. The standardized signalportion preferably has a data content which is known on the receiveside.

[0017] Based on the estimated channel coefficients, estimates for theinterference components have to be provided. To this end, in a firststep two or more estimated channel coefficients of a specific channelmay be processed exploiting a correlation among the two or moreestimated channel coefficients. By means of this processing aninterference of the individual channels can be simulated. Therefore, ina second step, estimates for the interference components, which arecomprised within the estimated channel coefficients and which wereartificially introduced, e.g. during the estimation step, can be derivedfrom the processed channel coefficients. Usually, step-like structureswithin the estimated channel coefficients of a specific channel arebroken as a result of the processing.

[0018] Processing the estimated channel coefficients by utilizing theircorrelation can be performed in various ways. It is preferably conductedsuch that the correlation among a plurality of estimated channelcoefficients belonging to different instants is exploited. As anexample, the estimated channel coefficients may undergo an interpolationor a filtering step. The interpolation may be effected by linearinterpolation. Filtering has the additional advantage that it not onlyexploits the correlation among the estimated channel coefficients of aspecific channel, but also brings about an efficient noise suppression.

[0019] Preferably, the processed channel coefficients are not only usedfor deriving estimates for the interference components, but also as abasis for determining improved estimates for the channel coefficients bytaking into account the estimated interference components. For example,determining the improved estimates may comprise subtracting theestimated interference components from the processed channelcoefficients. Thus, interference cancellation is effected.

[0020] The above method can be implemented both as a computer programproduct comprising program code portions for performing the method andas a hardware solution. The hardware solution is constituted by anappropriately configured estimating circuit for estimating channelcoefficients in a multi carrier transmit diversity system operating inaccordance with a block-coding scheme. The estimating circuit comprisesa unit for determining from a receive signal for each channel estimatedchannel coefficients comprising artificially introduced interferencecomponents from adjacent channels, and a unit for deriving estimates forthe interference components and for determining interference-compensatedestimates for the channel coefficients utilizing the estimates for theinterference components. Preferably, the estimating circuit furthercomprises a processing unit for processing two or more of the estimatedchannel coefficients such that a correlation among the estimated channelcoefficients is exploited.

[0021] According to a preferred embodiment, it is determined for one ormore of the data blocks in dependence on at least one transmissionconstraint if the data words of said one or more data blocks are to bemultiplexed in the time domain or in the frequency domain. The datawords of the data blocks may thereafter be multiplexed in accordancewith the result of the determination. It may thus be decided on a datablock level how the data words are to be multiplexed. The decision onthe data block level allows to change the multiplexing domain from onedata block to a subsequent data block which is advantageous if one hasto cope with specific predefined or varying transmission constraints.Such a multiplexing method can be applied in various wirelesscommunication systems without major changes due to the specificmultiplexing flexibility gained by selecting the multiplexing domain onthe data block level. The multiplexing domain can be determined for eachdata block individually or simultaneously for a plurality of datablocks. For example, it can be decided for a sequence of data blocksthat all data words comprised within the sequence of data blocks are tobe multiplexed in either the time domain or in the frequency domain.

[0022] The multiplexing domain is preferably determined by taking intoaccount one or more transmission constraints. For example, thetransmission constraints may comprise one or more physical transmissionconstraints or one or more data-related transmission constraints. Thetransmission constraints can also comprise both one or more physicaltransmission constraints as well as one or more data-relatedtransmission constraints. The physical transmission constraints relateto the physical transmission conditions and can be derived from physicaltransmission parameters like a channel coherence bandwidth or acoherence time. The data-related transmission constraints relate tosystem specific constraints regarding for example the employedmultiplexing scheme for the data words, a given structure of the datasignal, a given structure of the data blocks, a given structure of thedata words or a given structure of the data symbols.

[0023] The physical transmission constraints may be determined based onat least one of a channel coherence bandwidth

B _(C)≈1/τ_(rms)   (1)

[0024] and a coherence time

t _(C)≈1/(2·f _(D)),   (2)

[0025] wherein f_(D) is the doppler frequency and τ_(rms) is the rootmean square of the delay spread of the channel impulse response.

[0026] Many transmit diversity schemes require constant or at leastapproximately constant channel parameters during transmission of onedata word. If the data words are to be multiplexed in the frequencydomain, a comparatively large coherence band width is required. Thismeans that the relation

B _(C) >>N/T   (3)

[0027] has to be fulfilled at least approximately, wherein N is thenumber of data symbols per data word and T is the duration of one of thedata symbols, i.e. the duration of one time slot. A comparatively largecoherence bandwidth requires that the channel coefficients of N adjacentsubcarriers have to be almost constant.

[0028] On the other hand, if the data words are to be multiplexed in thetime domain, a comparatively large coherence time is required. Thismeans that the relation

t _(C) >>T·N   (4)

[0029] has to be fulfilled at least approximately. In other words: Nsubsequent data symbols have to have nearly constant channel parameters,i.e. the channel coefficients for a single subcarrier have to remainconstant for a period of N·T.

[0030] The physical transmission constraint may be determined byassessing if one or both of the relations (3) and (4) are fulfilled.Dependent on which of the two relations (3) and (4) is fulfilled best,it may be decided that the data words of the data blocks are to bemultiplexed either in the time domain or in the frequency domain as ageneral rule. Deviations from this general rule may become necessary dueto data-related transmission constraints.

[0031] The data symbols may be derived from the data signal in variousways dependent on the block-coding scheme which is used. If, forexample, the block-coding scheme of permutation in the time or frequencydomain is used, the data symbols contained in the data words arepermutations of information symbols comprised within the data signal. Asa further example, if the block-coding scheme of space-time orspace-frequency block-coding is used, the data symbols contained in thedata words are obtained from the information symbols comprised withinthe data signal by means of permutation and basic arithmetic operations,such as negation and complex conjugation.

[0032] The data signal from which the one or more data blocks aregenerated can have any format. According to a preferred embodiment, thedata signal has the format of a sequence of discrete informationsymbols. For example, the data signal may have the structure of vectors,each vector comprising a predefined number of information symbols. Thenature of the information symbols may depend on the specific wirelesscommunication system in which the method according to the invention isused. Many wireless communication systems employ different types ofinformation symbols for different purposes. For example, some wirelesscommunication systems use data signals which comprise a preamble, one ormore user data sections or both a preamble and one or more user datasections. Usually, the preamble has a predefined structure and enhancesfunctions like channel estimation, frequency synchronization and timingsynchronization.

BRIEF DESCRIPTION OF THE DRAWINGS

[0033] Further advantages of the invention will become apparent byreference to the following description of preferred embodiments of theinvention in the light of the accompanying drawings, in which:

[0034]FIG. 1 shows a data signal in the form of a physical burst to beprocessed in accordance with the invention;

[0035]FIG. 2 shows the structure of an OFDM symbol comprising a cyclicprefix;

[0036]FIG. 3 is a block diagram of a transmitter stage of a transceiverfor wireless communication;

[0037]FIG. 4 shows several modulation schemes defined in the HIPERLAN/2standard;

[0038]FIG. 5 shows a block code encoder of the transceiver depicted inFIG. 3;

[0039]FIG. 6 shows a configuration of a transmit antenna diversityscheme;

[0040]FIG. 7 is a schematic diagram of multiplexing data words in thetime domain;

[0041]FIG. 8 is a schematic diagram of multiplexing data words in thefrequency domain.

[0042]FIG. 9 is a block diagram of a receiver stage of a transceiver forwireless communication;

[0043]FIG. 10 shows a more detailed block diagram of the receiver stageof FIG. 9; and

[0044]FIG. 11 is a graphical representation of the estimated channelcoefficients prior to and after a correlation operation.

DESCRIPTION OF PREFERRED EMBODIMENTS

[0045] Although the present invention can be used in any multi carriertransmit diversity system which employs a transmit diversity schemeallowing to generate data blocks having a structure similar to e.g. aSTBC code matrix, the following description of preferred embodiments isexemplarily set forth with respect to a multi carrier system whichemploys orthogonal frequency division multiplexing (OFDM) and whichalternately utilizes STBC and SFBC for generating data blocks from adata signal.

[0046] The exemplary multi carrier system is derived from the Europeanwireless local area network (WLAN) standard HIgh PErformance Radio LocalArea Network type 2 (HIPERLAN/2). HIPERLAN/2 systems are intended to beoperated in the 5 GHz frequency band. Up to now, the HIPERLAN/2 systemand many other wireless communications systems do not support transmitdiversity in spite of the fact that transmit diversity would improve thetransmission performance and reduce negative effects of fast fading likeRayleigh fading. A system overview of HIPERLAN/2 is given in ETSI TR 101683, Broadband Radio Access Networks (BRAN); HIPERLAN Type 2; SystemOverview, V1.1.1 (2000-02) and the physical layer of HIPERLAN/2 isdescribed in ETSI TS 101 475; Broadband Radio Access networks (BRAN);HIPERLAN Type 2; Physical (PHY) Layer, V1.1.1 (2000-04). The multicarrier scheme of OFDM, which is specified in the HIPERLAN/2 standard,is very robust in frequency selective environments.

[0047] In FIG. 1 a typical physical burst of HIPERLAN/2 is illustrated.The physical burst comprises a preamble consisting of preamble symbolsand a user data section consisting of user data symbols. In HIPERLAN/2five different physical bursts are specified. Three of the five physicalbursts have a different preamble each and the two remaining physicalbursts have a further preamble in common. The last three preamblesymbols constitute a periodic structure which is identical for allpreamble types. This periodic structure consists of a short OFDM symbolC32 of 32 samples followed by two identical regular OFDM symbols C64 of64 samples. The short OFDM symbol C32 is a cyclic prefix which is arepetition of the second half of one of the C64 OFDM symbols. Theso-called C-preamble depicted in FIG. 1 is used in HIPERLAN/2 forchannel estimation, frequency synchronization and timingsynchronization. The periodic structure within the C-preamble isnecessary in order to allow the use of synchronization algorithms withcomparatively low complexity.

[0048] The user data section of the physical burst depicted in FIG. 1comprises a variable number N_(SYM) of OFDM symbols required to transmita specific protocol data unit (PDU) train. Each OFDM symbol of the userdata section consists of a cyclic prefix and a useful data part. Thecyclic prefix consists of a cyclic continuation of the useful data partand is inserted before it. Thus, the cyclic prefix is a copy of the lastsamples of the useful data part as depicted in FIG. 2.

[0049] The length of the useful data part of the physical burst shown inFIG. 1 is equal to 64 samples and has a duration of 3.2 μs. The cyclicprefix has a length of either 16 (mandatory) or 8 (optional) samples anda duration of 0.8 μs or 0.4 μs, respectively. Altogether, a OFDM symbolshas a length of either 80 or 72 samples corresponding to a symbolduration of 4.0 μs or 3.6 μs, respectively. An OFDM symbol therefore hasan extension in the time domain. A OFDM symbol further has an extensionin the frequency domain. According to HIPERLAN/2, a OFDM symbol extendsover 52 subcarriers. 48 subcarriers are reserved for complex valuedsubcarrier modulation symbols and 4 subcarriers are reserved for pilots.

[0050] For typical HIPERLAN/2 scenarios the above relation (4) isusually fulfilled because the doppler frequency f_(D) is comparativelylow. However, especially in outdoor environments, relatively large delayspreads can occur. Consequently, relation (3) cannot always befulfilled. Therefore, a transmit diversity scheme like STBC multiplexingin the time domain should generally be a preferred transmit diversityscheme for a HIPERLAN/2 scenario from the point of view that the channelover one space-time data word should be as constant as possible.However, the following and further problems may arise when STBC isapplied to physical bursts having the structure depicted in FIG. 1 or asimilar structure.

[0051] Both the physical burst and the OFDM symbols comprised thereinhave predefined dimensions in the time domain and in the frequencydomain. Concurrently, STBC requires that each STBC data word has apredetermined length N. Thus, data unit fitting problems arise if thedimension of e.g. an OFDM symbol of the preamble or of the user datasection cannot be mapped on an integer multiple of the length of oneSTBC data word. Moreover, when applying STBC to the periodic C-preambledepicted in FIG. 1, the periodicity of the C-preamble gets lost. This isdue to the fact that the one or more STBC data words relating to thesecond C64 OFDM symbol will no longer be equal to the one or more STBCdata words relating to the first C64 OFDM symbol. The loss ofperiodicity, however, leads to the problem that the symbolsynchronization algorithms which make use of a periodic structure withinthe preamble can no longer be employed. Also, the C32 OFDM symbol cannotserve any longer as a guard interval separating the OFDM symbols withinthe preamble. The reason therefore is that in case of multipathpropagation the first C64 OFDM symbol interferes with the second C64OFDM symbol which is no longer equal to the first C64 OFDM symbol. Theabove problems and further problems not explicitly discussed above donot occur when the data words are multiplexed in the frequency domain.Therefore it is advantageous to dynamically switch between STBC and SFBCon a data block level during transmission.

[0052] In FIG. 3, the physical layer of a transmitter stage 10 of atransceiver is illustrated. The transmitter stage 10 comprises ascrambler 12, an FEC coding unit 14, an interleaving unit 16, a mappingunit 18, an OFDM unit 20, a burst forming unit 22, a block code encoder24, a multiplexer 26, a radio transmitter 30 and a control unit 32. Theblock code encoder 24 and the multiplexer 26 together form anencoder/multiplexer unit 28.

[0053] The transmitter stage 10 depicted in FIG. 3 receives as inputsignal a PDU train from a data link control (DLC). Each PDU trainconsists of information bits which are to be framed into a physicalburst, i.e. a sequence of OFDM symbols to be encoded, multiplexed andtransmitted.

[0054] Upon receipt of a PDU train the transmission bit rate within thetransceiver is configured by choosing an appropriate physical mode basedon a link adaption mechanism. A physical mode is characterized by aspecific modulation scheme and a specific code rate. In the HIPERLAN/2standard several different coherent modulation schemes like BPSK, QPSK,16-QAM and optional 64-QAM are specified. Also, for forward errorcontrol, convolutional codes with code rates of ½, {fraction (9/16)} and¾ are specified which are obtained by puncturing of a convolutionalmother code of rate ½. The possible resulting physical modes aredepicted in FIG. 4. The data rate ranging from 6 to 54 Mbit/s can bevaried by using various signal alphabets for modulating the OFDMsubcarriers and by applying different puncturing patterns to a motherconvolutional code.

[0055] Once an appropriate physical mode has been chosen, the NBPDUinformation bits contained within the PDU train are scrambled with thelength-127 scrambler 12. The scrambled bits are then output to the FECcoding unit 14 which encodes the NBPDU scrambled PDU bits according tothe previously set forward error correction.

[0056] The encoded bits output by the FEC coding unit 14 are input intothe interleaving unit 16 which interleaves the encoded bits by using theappropriate interleaving scheme for the selected physical mode. Theinterleaved bits are input into the mapping unit 18 where sub-carriermodulation is performed by mapping the interleaved bits into modulationconstellation points in accordance with the chosen physical mode. Asmentioned above, the OFDM subcarriers are modulated by using BPSK, QPSK,16-QAM or 64-QAM modulation depending on the physical mode selected fordata transmission.

[0057] The mapping unit 18 outputs a stream of complex valued subcarriermodulation symbols which are divided in the OFDM unit in groups of 48complex numbers. In the OFDM unit a complex base band signal is producedby OFDM modulation as described in ETSI TS 101 475, Broadband RadioAccess Networks (BRAN); HIPERLAN Type 2; Physical (PHY) Layer, V1.1.1(2000-04).

[0058] The complex base band OFDM symbols generated within the OFDM unit20, where pilot subcarriers are inserted, are input into the physicalburst unit 22, where an appropriate preamble is appended to the PDUtrain and the physical burst is built. The physical burst produced bythe physical burst unit 22 has a format as depicted in FIG. 1. Thephysical burst unit 22 thus outputs a sequence of complex base band OFDMsymbols in the form of the physical burst to the block code encoder 24.

[0059] The function of the block code encoder 24 is now generallydescribed with reference to FIG. 5. In general, the block code encoder24 receives an input signal in the form of a sequence of vectors X=[X₁X₂. . . X_(K)]^(T) of the length K. The block code encoder 24 encodes eachvector X and outputs for each vector X a data block comprising aplurality of signal vectors C⁽¹⁾, C⁽²⁾ . . . , C^((M)) as depicted inFIG. 5. Each signal vector C⁽¹⁾, C⁽²⁾ . . . , C^((M)) corresponds to asingle data word. Thus, the data block generated from the vector Xcomprises M data words, wherein M is the number of transmitter antennas.

[0060] Each data word C^((i)) with i=1 . . . M comprises N data symbols,i.e. each data word C^((i)) has a length of N. The value of N cannot befreely chosen since the matrix C spanned by the data words C^((i)) hasto be orthogonal in this embodiment. Several examples for data blocks inthe form of orthogonal code matrices C are described in U.S. Pat. No.6,088,408 herewith incorporated by reference. In the block-codingapproach described in the present embodiment, all data symbols c_(j)^(i) of the code matrix C are derived from the components of the inputvector X and are simple linear functions thereof or of its complexconjugate.

[0061] If a receive signal vector Y at one receive antenna is denoted byY=[Y₁Y₂. . . Y_(N)]^(T), the relationship between Y and the code matrixC is as follows: $\begin{matrix}{\begin{bmatrix}Y_{1} \\Y_{2} \\\ldots \\Y_{N}\end{bmatrix} = {\begin{bmatrix}c_{1}^{(1)} & c_{1}^{(2)} & \ldots & c_{1}^{(M)} \\c_{2}^{(1)} & \ldots & \quad & c_{2}^{(M)} \\\ldots & \quad & \ldots & \ldots \\c_{N}^{(1)} & c_{N}^{(2)} & \ldots & c_{N}^{(M)}\end{bmatrix} \cdot \begin{bmatrix}h^{(1)} \\h^{(2)} \\\ldots \\h^{(M)}\end{bmatrix}}} & (5)\end{matrix}$

[0062] where h^((i)) represents the channel coefficient of the channelfrom the i-th transmit antenna to the receive antenna. A generalizationto more receive antennas is straightforward.

[0063] In the following, examples of possible block code matrices fortwo and three transmitter antennas, respectively, are discussed in moredetail. The configuration of a wireless communication system with twotransmit antennas and one receive antenna is depicted in FIG. 6. Thewireless communications system of FIG. 6 comprises two transmitchannels, each transmit channel being characterized by a specificchannel coefficient h^((i)) with i=1,2.

[0064] For two transmit antennas one possible block code matrix C with acode rate R=1 is $\begin{matrix}{C = \begin{bmatrix}X_{1} & X_{2} \\{- X_{2}^{*}} & X_{2}^{*}\end{bmatrix}} & (6)\end{matrix}$

[0065] For three transmit antennas one possible block code matrix C witha code rate R=0,5 is: $\begin{matrix}{C = \begin{bmatrix}X_{1} & X_{2} & X_{3} \\{- X_{2}} & X_{1} & {- X_{4}} \\{- X_{3}} & X_{4} & X_{1} \\{- X_{4}} & {- X_{3}} & X_{2} \\X_{1}^{*} & X_{2}^{*} & X_{3}^{*} \\{- X_{2}^{*}} & X_{1}^{*} & {- X_{4}^{*}} \\{- X_{3}^{*}} & X_{4}^{*} & X_{1}^{*} \\{- X_{4}^{*}} & {- X_{3}^{*}} & X_{2}^{*}\end{bmatrix}} & (7)\end{matrix}$

[0066] The code rate R is defined as the ratio of the length K of theinput vector X and the length N of each code word C^((i)):

R=K/N   (8)

[0067] As can be seen from FIG. 5, the block code encoder 24 outputs foreach data signal in the form of a vector X a data block in the form of amatrix C. The data block output by the block code encoder 24 is inputinto the multiplexer 26 which multiplexes the data words (vectorsC^((i))) of each datablock in accordance with an externally providedcontrol signal either in the time domain or in the frequency domain. Thecontrol signal is generated by the control unit 32 based on anassessment of the transmission constraints.

[0068] In the multi carrier scheme OFDM, the output of the block codeencoder 24 is modulated onto subcarriers which are orthogonal to eachother. There exist essentially two possibilities to multiplex a datablock comprising individual data words in an OFDM system. According to afirst possibility depicted in FIG. 7, the data words of a specific datablock are extended in the time direction (STBC). In other words: Thedata words are multiplexed in the time domain. According to a secondpossibility, the data words of a data block are extended in thefrequency direction as depicted in FIG. 8 (SFBC).

[0069] As can be seen from FIGS. 7 and 8, the individual data words of adata block are transmitted from different transmit antennas. Accordingto the multiplexing scheme of FIG. 7, an individual data block istransmitted on an individual subcarrier over a time interval of N·T,wherein N is the number of data symbols per data word and T is theduration of one of the data symbols. According to the multiplexingscheme of FIG. 8, an individual data block is spread over N subcarriersand is transmitted during a time interval of T. It can clearly be seenthat the multiplexing scheme of FIG. 7 can generally be employed whenrelation (4) is fulfilled, whereas the multiplexing scheme of FIG. 8 cangenerally be employed when relation (3) is satisfied.

[0070] The encoded and multiplexed output signal of theencoder/multiplexer unit 28 is input into the radio transmitter 30. Theradio transmitter 30 performs radio transmission over a plurality oftransmit antennas by modulating a radio frequency carrier with theoutput signal of the encoder/multiplexer unit 28.

[0071] The transceiver with the transmitter stage 10 of FIG. 3 furthercomprises a receiver stage not depicted in FIG. 3. The receiver stagehas a physical layer with components for performing the inverseoperations of the components depicted in FIG. 3. For example, thereceiver stage comprises a descrambler, a FEC decoding unit, ademultiplexer/decoder unit with a demultiplexer and a block codedecoder, etc.

[0072]FIG. 9 shows some components of such a receiver stage 40 of thetransceiver with the transmitter stage 10 depicted in FIG. 3. As becomesapparent from FIG. 9, a receive signal X vector Y received via a receiveantenna 42 is concurrently fed to a channel estimation circuit 44 and ademodulator 46. In the following, operation of the channel estimatingcircuit 44 is exemplarily described for the case that two transmitantennas and one receive antenna 42 are utilized (FIG. 6).

[0073] In this case encoding of the data signal can be performed on thebasis of the above block code matrix (6) and the receive signal vectorcan be written as Y=[Y_(j)Y_(j+1)]^(T). The index j denotes a specificinstant in time or frequency depending on whether STBC or SFBC isutilized.

[0074] At a first instant j, X_(i) is transmitted from the firsttransmit antenna and X_(i+1) is transmitted from the second transmitantenna. At a subsequent instant j+1, −X*_(i+1) is transmitted from thefirst transmit antenna and X*_(I) is transmitted from the secondtransmit antenna. The individual components Y_(j) and Y_(j+1) of thereceive signal vector Y can thus be written as

Y _(j) =X _(i) ·h ⁽¹⁾(z _(j))+X _(i+1) ·h ⁽²⁾(z _(j))+n _(j)

Y _(j+1) =−X* _(i+1) ·h ⁽¹⁾(z _(j+1))+X* _(i) ·h ⁽²⁾(z _(j+1))+n _(j+1)  (9)

[0075] The instant variable z_(j) denotes either the time index t_(j) ifSTBC is applied or the frequency index f_(j) if SFBC is applied.Therefore, h^((i))(z_(j)) is the coefficient of the channel between thetransmit antenna i and the receive antenna for a data symbol transmittedat t_(j) (STBC) or via the frequency f_(j) (SFBC). The term n_(j)denotes the white gaussian noise at the instant z_(j).

[0076] For the case that STBC is applied (z_(j)=t_(j)) and the coherencetime tc is relatively large, i.e. if relation (4) is fulfilled, or forthe case that SFBC is applied (z_(j)=f_(j)) and the coherence bandwidthB_(C) is relatively large, i.e. if relation (3) is fulfilled, thefollowing assumptions are valid

h ⁽¹⁾(z _(j))=h ⁽¹⁾(z _(j+1))=h ⁽¹⁾

h ⁽²⁾(z _(j))=h ⁽²⁾(z _(j+1))=h ⁽²⁾   (10)

[0077] This means that if either the coherence time t_(C) or thecoherence bandwidth B_(C) is relatively large, and if additionally theappropriate block-coding scheme was chosen, equations (9) become

Y _(j) =X _(i) ·h ⁽¹⁾ +X _(i+1) ·h ⁽²⁾ +n _(j)

Y _(j+1) =−X* _(i+1) ·h ⁽¹⁾ +X* _(i) ·h ⁽²⁾ +n _(j+1)   (11)

[0078] Equation (11) can be written in terms of the receive signalvector Y and a data matrix Z, which is equivalent to the code matrix C,as $\begin{matrix}{Y = {\begin{bmatrix}Y_{j} \\Y_{j + 1}\end{bmatrix} = {{{\begin{bmatrix}X_{i} & X_{i + 1} \\{- X_{i + 1}^{*}} & X_{i}^{*}\end{bmatrix} \cdot \begin{bmatrix}h^{(1)} \\h^{(2)}\end{bmatrix}} + \begin{bmatrix}n_{j} \\n_{j + 1}\end{bmatrix}} = {{Z \cdot H} + N}}}} & (12)\end{matrix}$

[0079] In order to provide an estimate for the channel coefficients h⁽¹⁾and h⁽²⁾, the receive signal vector Y is multiplied with the HermitianZ^(H) of the known data matrix Z. The content of the known data matrix Zcorresponds to a standardized preamble portion which is known to thetransceiver. The multiplication of Z^(H) and Y yields

Z ^(H) ·Y=Z ^(H) Z·H+Z ^(H) ·N=Ĥ  (13)

[0080] Since Z is a scaled unitary matrix, i.e. $\begin{matrix}{Z^{- 1} = {\frac{1}{\det (X)}Z^{H}}} & (14)\end{matrix}$

[0081] the channel coefficients in equation (13) are separated.

[0082] The estimated channel coefficients thus obtained were derivedbased on the assumption that the individual channels do not changeduring an amount of instants z_(j) required to transmit one data word.However, in many cases equations (11) are not valid. When applying STBC,this means that the channel changes during the amount of instantsrequired to transmit one space-time data word, i.e. during N adjacentdata symbols, because the coherence time t_(C) is comparatively small.In the case of two transmit antennas this means thath^((i))(t)≠h^((i))(t+T), where i=1,2 and where T denotes the duration ofa data symbol. When applying SFBC, the problem arises that the channelchanges over N adjacent subcarriers, which are required to transmit onespace-frequency code word, because the coherence bandwidth B_(C) iscomparatively small. In other words, h^((i))(f)≠h^((i))(f+Δf), wherei=1,2 and where Δf=1/T denotes the subcarrier spacing. As a consequenceof a fact that equations (3) and (4), and thus equations (10), aregenerally not valid, equation (13) will even in a noiseless case notyield the exact channel coefficients.

[0083] In order to improve the estimates for the channel coefficients,the estimating circuit 44 of the receiver stage 40 depicted in FIG. 9 isconstructed as depicted in FIG. 10. As becomes apparent from FIG. 10,the estimating circuit 44 according to the invention comprises a unit 48for determining from a receive signal vector Y estimated channelcoefficients for each of the channels, the estimated channelcoefficients comprising artificially introduced interference componentsfrom adjacent channels, a processing unit 50 for exploiting acorrelation among the estimated channel coefficients in order to obtainprocessed channel coefficients, and a unit 52 for deriving estimates forthe interference components and for determining interference-compensatedestimates for the channel coefficients using the estimates for theinterference components.

[0084] In the following, the operation of the channel estimating circuit44 depicted in FIG. 9 will be described in more detail.

[0085] In a first step, the unit 48 for determining the estimatedchannel coefficients receives the receive signal vector Y. Within theunit 48, the receive signal vector Y is multiplied with the HermitianZ^(H) of the known data matrix Z corresponding to the standardized datasymbols comprised within a preamble portion of the transmitted OFDM datasignal. One thus obtains

Z ^(H) ·Y=Ĥ  (15)

[0086] Since Z^(H) can be written as $\begin{matrix}{Z^{H} = \begin{bmatrix}X_{i}^{*} & {- X_{i + 1}} \\X_{i + 1}^{*} & X_{i}\end{bmatrix}} & (16)\end{matrix}$

[0087] and since Ĥ can be written as Ĥ[ĥ⁽¹⁾(z)ĥ⁽²⁾(z)]^(T), equation(15) can be written as

ĥ ⁽¹⁾(z _(j))=ĥ ⁽¹⁾(z _(j+1))=X* _(i) ·Y _(j) −X _(i+1) ·Y _(j+1)

ĥ ⁽²⁾(z _(j))=ĥ ⁽²⁾(z _(j+1))=X* _(i+1) ·Y _(j) +X _(i) ·Y _(j+1)   (17)

[0088] The above equation (17) departs from the assumption that

h ^((i))(z _(j))=h ⁽¹⁾(z _(j+1))=h ^((i))(z)   (18)

[0089] This assumption corresponds to equation (10) and was made becauseit is impossible to obtain the four unknown parameters h⁽¹⁾(z_(j)),h⁽¹⁾(z_(j+1)), h⁽²⁾(z_(j)) and h⁽²⁾(z_(j+1)) from the two only equations(9). In other words, equations (9) are under-determined.

[0090] Now the values of Y_(j) ad Y_(j+1) of equations (9) are insertedinto equations (17) and one obtains:

ĥ ⁽¹⁾(z)=|X _(i)|² ·h ⁽¹⁾(z _(j))+|X _(i+1)|² ·h ^((i))(z _(j+1))+X*_(i) ·X _(i+1) ·{h ⁽²⁾(z _(j))−h ⁽²⁾(z _(j+1))}+X* _(i) ·n _(j) −X_(i+1) ·n _(j+1)

ĥ ⁽²⁾(z)=|X _(i+1)|² ·h ⁽²⁾(z _(j))+|X _(i)|² ·h ⁽²⁾(z _(j+1))+X* _(i)·X _(i+1) ·{h ⁽¹⁾(z _(j))−h ⁽¹⁾(z _(j+1))}+X* _(i+1) ·n _(j) −X _(i) ·n_(j+1)   (19)

[0091] As becomes apparent from the above equation (19), the estimatedchannel coefficients ĥ⁽¹⁾(z) and ĥ⁽²⁾(z) comprise three differentcomponents, namely a useful component U^((i)), an interference componentI^((i)) and a noise component N^((i)). Equations (19) can therefore alsobe written as $\begin{matrix}\begin{matrix}{{{\hat{h}}^{(1)}(z)} = {{U^{(1)}\left( {{h^{(1)}\left( z_{j} \right)},{h^{(1)}\left( z_{j + 1} \right)}} \right)} +}} \\{{{I^{(1)}\left( {{h^{(2)}\left( z_{j} \right)},{h^{(2)}\left( z_{j + 1} \right)}} \right)} + N^{(1)}}} \\{= {U^{(1)} + I^{(1)} + N^{(1)}}}\end{matrix} & (20) \\\begin{matrix}{{{\hat{h}}^{(2)}(z)} = {{U^{(2)}\left( {{h^{(2)}\left( z_{j} \right)},{h^{(2)}\left( z_{j + 1} \right)}} \right)} +}} \\{{{I^{(2)}\left( {{h^{(1)}\left( z_{j} \right)},{h^{(1)}\left( z_{j + 1} \right)}} \right)} + N^{(2)}}} \\{{= {U^{(2)} + I^{(2)} + N^{(2)}}},}\end{matrix} & (21)\end{matrix}$

[0092] whereby

I ⁽¹⁾ =X* _(i) X _(i+1) ·{h ⁽²⁾(z _(j))−h ⁽²⁾(z _(j+1)})

I ⁽²⁾ =X _(i) ·X* _(i+1) {h ⁽¹⁾(z _(j))−h ⁽¹⁾(z _(j+1))}  (22)

[0093] The interference components I^((i)) of the estimated channelcoefficients ĥ^((i))(z) are a manifestation of block-coding, of theassumptions (10) and (18) that the channels do not change during anamount of instants required to transmit one data word, and of the factthat the channels do change over N instants, i.e. the fact thatassumptions (10) and (18) are not valid. The interference components arethus not present in the actual channel coefficients. Therefore, acancellation of these interference components I^((i)) comprised withinthe estimated channel coefficients ĥ^((i)) would lead to improvedestimates for the channel coefficients h^((i)).

[0094] Due to the fact that equation (9) is under-determined, it is notpossible to determine the interference components I^(i) exactly.Instead, estimates Î^((i)) have to be derived. In the following, apossible approach for deriving estimates Î^((i)) for the interferencecomponents I^((i)) is exemplarily described with reference to FIG. 11.

[0095] In FIG. 11, the crosses denote the estimated channel coefficientsĥ^((i)) of channel i for a series of instants z_(j). These estimatedchannel coefficients ĥ^((i)) were determined by the unit 48 forproviding estimated channel coefficients as described above. From FIG.11 it can clearly be seen that the estimated channel coefficientsĥ^((i)) of two neighboring instants z_(j) and z_(j+1) are identical.This is an expression of the assumption according to equations (10) and(18). However, since two neighboring estimated channel coefficientsĥ^((i)) are identical, the estimated channel coefficients ĥ^((i)) cannotbe used as a basis for actually calculating the interference components.This becomes apparent from equation (22) because if the appropriateestimated channel coefficients h^((i)) were inserted, the interferencecomponents I^((i)) would all be zero.

[0096] It is therefore necessary to break the identity of estimatedchannel coefficients ĥ^((i)) belonging to adjacent instants in order toobtain more realistic estimates for the actual channel coefficientsh^((i)). One way of breaking the identity among estimated channelcoefficients ĥ^((i)) belonging to adjacent instants are methods likefiltering which exploit the correlation among the estimated channelcoefficient ĥ^((i)). To that end, the estimation circuit 44 of thereceiver stage 40 depicted in FIG. 10 comprises a processing unit 50with a plurality of filters 50 ₁, 50 ₂ . . . 50 _(M).

[0097] For the filtering process within a specific filter 50 _(i), allestimated channel coefficients h^((i))(z_(j)) for the channel i are used(j=1 . . . N_(X), N_(X) denotes the number of subcarriers in thefrequency domain or the number of data symbols in the time domain towhich the filtering process should be applied). If SFBC is applied forexample, the filtering process may be applied to all used subcarriers.

[0098] If a low-pass filter g(z) is used for the filtering process, thefiltered estimated channel coefficients ĥ_(F) ^((i)) output by theprocessing unit 50 can be written as $\begin{matrix}{{{\hat{h}}_{F}^{(i)}(z)} = {{{{\hat{h}}^{(1)}(z)}*{g(z)}} = {\sum\limits_{m = 1}^{N_{x}}{{{\hat{h}}^{(i)}(m)} \cdot {g\left( {z - m} \right)}}}}} & (23)\end{matrix}$

[0099] During the filtering process, different estimates for the channelcoefficients h^((i))(z_(j)) and h^((i))(z_(j+1)) are obtained becausethe filtering exploits the correlation among the individual estimatedchannel coefficients ĥ^((i)) of adjacent instants. Of course, thefiltering process is only one possibility if exploiting thiscorrelation. Other processes like linear interpolation could also beused. However, the filtering process is especially advantageous becauseit not only provides the required different estimates for the channelcoefficients of adjacent instants, but also suppresses the noisecomponents N^((i)) present in the estimated channel coefficientsĥ^((i)). As becomes apparent from equations (20) and (21), the processed(i.e. filtered) estimated channel coefficients ĥ_(F) ^((i))(z) can bewritten as

ĥ _(F) ^((i))(z)=U _(F) ^((i)) +I _(F) ^((i)) +N _(F) ^((i))   (24)

with

I _(F) ⁽¹⁾ =X* _(i) ·X _(i+1) ·{h _(F) ⁽²⁾(z _(j))−h _(F) ⁽²⁾(z _(j+1))}

I _(F) ⁽²⁾ =X* _(i) ·X _(i+1) ·{h _(F) ⁽¹⁾(z _(j))−h _(F) ⁽¹⁾(z_(j+1))}  (25)

[0100] Estimates Î_(F) ⁽¹⁾ for the interference components I_(F) ^((i))can be calculated on the basis of the processed channel coefficientsĥ_(F) ^((i))(z). Equations (25) then yield

Î _(F) ⁽¹⁾ =X* _(i) ·X _(i+1) ·{ĥ _(F) ⁽²⁾(z _(j))−ĥ _(F) ⁽²⁾(z _(j+1))}

Î _(F) ⁽²⁾ =X* _(i) ·X _(i+1) ·{ĥ _(F) ⁽¹⁾(z _(j))−ĥ _(F) ⁽¹⁾(z_(j+1))}  (26)

[0101] These calculations are performed within the unit 52 of theestimation circuit 44 illustrated in FIG. 10. The calculation takes intoaccount the processed channel coefficients ĥ_(F) ^((i)) received fromthe correlation unit 50.

[0102] The estimates Î_(F) ^((i)) for the interference components Î_(F)^((i)) are used for interference cancellation. Thus, improved estimatesĥ_(F+IC) ^((i))(z) are determined within the unit 52 as follows

ĥ _(F+IC) ^((i))(z)=ĥ _(F) ^((i))(z)−Î _(F) ^((i))   (27)

[0103] After the improved estimates ĥ_(F+IC) ^((i)) have beendetermined, they are passed from the unit 52 in the form of a matrixĤ_(F+IC) to the demodulator 46 of the receiver stage 40.

[0104] The basic concept underlying the invention can be extended totransmit diversity systems comprising more than two transmit antennas.The following embodiment of the invention is based on a transmitdiversity system comprising three transmit antennas and operating inaccordance with a block-coding scheme using the code matrix shown inequation (7). For the noiseless case, equation (11) can be written as

Y=Z·H   (28)

[0105] which is equivalent to $\begin{matrix}\begin{matrix}{Y_{j} = {{X_{i} \cdot h_{j}^{(1)}} + {X_{i + 1} \cdot h_{j}^{(2)}} + X_{i + 2} + h_{j}^{(3)}}} \\{Y_{j + 1} = {{{- X_{i + 1}} \cdot h_{j + 1}^{(1)}} + {X_{i} \cdot h_{j + 1}^{(2)}} - {X_{i + 3} \cdot h_{j + 7}^{(3)}}}} \\{\ldots = \ldots} \\{Y_{j + 7} = {{{- X_{i + 3}^{*}} \cdot h_{j + 7}^{(1)}} - {X_{i + 2}^{*} \cdot h_{j + 7}^{(2)}} + {X_{i + 3}^{*} \cdot h_{j + 7}^{(3)}}}}\end{matrix} & (29)\end{matrix}$

[0106] In equation (29), the abbrevation ĥ^((i))(z_(j))=ĥ_(j) ^((i)) isused.

[0107] Departing from the equation

Z ^(H) ·Y=Z ^(H) ·Z·H   (30)

[0108] the channel coefficients h⁽¹⁾ of the first transmit antenna canbe derived using equation (29) follows $\begin{matrix}\begin{matrix}{{\hat{h}}_{j}^{(1)} = {\hat{h}}_{j + 1}^{(1)}} \\{= \ldots} \\{= {\hat{h}}_{j + 7}^{(1)}} \\{= {{X_{i}^{*}\left( {{X_{i} \cdot h_{j}^{(1)}} + {X_{i + 1} \cdot h_{j}^{(2)}} + {X_{i + 2} \cdot h_{j}^{(3)}}} \right)} -}} \\{{{X_{i + 1}^{*}\left( {{{- X_{i + 1}} \cdot h_{j + 1}^{(1)}} + {X_{i} \cdot h_{j + 1}^{(2)}} - {X_{i + 3} \cdot h_{j + 1}^{(3)}}} \right)} +}} \\{{- {X_{i + 3}\left( {{{- X_{i + 3}^{*}} \cdot h_{j + 7}^{(1)}} - {X_{i + 2}^{*} \cdot h_{j + 7}^{(2)}} + {X_{i + 3}^{*} \cdot h_{j + 7}^{(3)}}} \right)}}}\end{matrix} & (31)\end{matrix}$

[0109] From equation (31) the interference components can be deduced.The interference components for the first antenna are for example

X*_(i)X_(i+1)·h_(j) ⁽²⁾−X*_(i+1) X _(i)·h_(j+1) ⁽²⁾,   (32)

[0110] which is part of the interference from the second antenna, and

X*_(i)X_(i+2)·h_(j) ⁽³⁾−X*_(i+1) X _(i+3)·h_(j+1) ⁽³⁾,   (33)

[0111] which is part of the interference from the third transmitantenna.

[0112] The individual interference components are determined bysubjecting equation (31) to a filtering process as described earlier.Thereafter, interference cancellation is performed as outlined above toderive improved estimates for the channel coefficients.

1. A method of estimating channel coefficients (h) in a multi carriertransmit diversity system operating in accordance with a block-codingscheme, comprising: a) determining from a receive signal (Y) for eachchannel estimated channel coefficients (ĥ) comprising artificiallyintroduced interference components (I) from adjacent channels; b)deriving estimates (Î) for the interference components (I); and c)determining interference-compensated estimates (ĥ_(F+IC)) for thechannel coefficients (h) on the basis of the estimates (Î) for theinterference terms (I).
 2. The method of claim 1, wherein the estimatedchannel coefficients (h) are determined based on the assumption that thechannels do not change during an amount of instants (z) required totransmit two or more data symbols.
 3. The method according to claim 2,wherein, based on the assumption, the estimated channel coefficients (ĥ)are determined such that the estimated channel coefficients (ĥ) of twoor more adjacent instants (z) are identical.
 4. The method of one ofclaims 1 to 3, wherein determining the estimated channel coefficients(ĥ) comprises multiplying a known data matrix (Z) comprised within thereceive signal (Y) with the Hermitian (Z^(H)) of the known data matrix(Z).
 5. The method of one of claims 1 to 4, wherein the step ofdetermining estimates (Î) for the interference components (I) of aspecific channel comprises exploiting a correlation among a plurality ofchannel coefficients (ĥ) estimated for the specific channel.
 6. Themethod according to claim 5, wherein the estimated channel coefficients(ĥ) are processed such that for the specific channel an identity ofestimated channel coefficients (ĥ) which belong to adjacent instants (z)is broken.
 7. The method of claim 6, wherein processing of the estimatedchannel coefficients (ĥ) is effected by interpolation or filtering. 8.The method of one of claims 6 or 7, wherein the estimates (Î) for theinterference components (I) are derived from the processed channelcoefficients (ĥ_(F)).
 9. The method of one of claims 6 to 8 wherein theinterference-compensated estimates (ĥ_(F+IC)) for the channelcoefficients (h) are derived from the processed channel coefficients(ĥ_(F)).
 10. The method of claim 9, wherein determining theinterference-compensated estimates (ĥ_(F+IC)) comprises subtracting theestimates (Î) for the interference components (I) from the processedchannel coefficients (ĥ_(F)).
 11. The method of one of claims 1 to 10,wherein the block-coding is effected by space-time block-coding (STBC)or space-frequency block-coding (SFBC).
 12. The method of claim 11,further comprising switching between space-time block-coding (STBC) andspace-frequency block-coding (SFBC) in dependence on one or moretransmission constraints.
 13. A computer program product comprisingprogram code portions for performing the steps of one of claims 1 to 12when the product is run on a computer.
 14. The computer program productof claim 13 stored on a computer readable recording medium.
 15. Anestimating circuit (44) for estimating channel coefficients (h) in amulti carrier transmit diversity system operating in accordance with ablock-coding scheme, comprising: a) a unit (48) for determining from areceive signal (Y) for each channel estimated channel coefficients (ĥ)comprising artificially introduced interference components (I) fromadjacent channels; and b) a unit (52) for deriving estimates (Î) for theinterference components (I) and for determining interference-compensatedestimates (ĥ_(F+IC)) for the channel coefficients (h) on the basis ofthe estimates (Î) for the interference components (I).
 16. Theestimating circuit according to claim 15, further comprising aprocessing unit (50) for processing a plurality of channel coefficients(ĥ) estimated for a specific channel utilizing a correlation among theestimated channel coefficients (ĥ).
 17. A transceiver of a wirelesscommunication system comprising a receiver stage (40) with an estimatingcircuit (44) according to claim 15 or 16.